NO-TUNE SSB/CW TRANSCEIVER FOR 10GHz
Matjaz Vidmar, S53MV
1. Advantages and drawbacks of zero-IF transceivers
Zero-IF transceivers have both advantages and drawbacks
when compared to conventional SSB transceivers with crystal
filters and many frequency conversions. Considering the current
state of technology, zero-IF transceivers are probably most
suitable for the low amateur microwave bands: 1296MHz,
2304/2320MHz and 5760MHz. Therefore working radios for the
abovementioned frequency bands were developed first [1].
Although the published zero-IF transceivers for 1296MHz,
2304/2320MHz and 5760MHz allow many modifications and
improvements of the original design, one would like to extend
the zero-IF design to other frequency bands as well. However,
the transfer of a transceiver design to another frequency
band may not be straightforward.
Amateur-radio SSB transceivers for 432MHz and especially
144MHz require a very high dynamic range. A 144MHz SSB receiver
should both withstand local stations with kilowatt transmitters
as well as acheive a low noise figure. The dynamic range
requirement for a 144MHz SSB receiver is probably even more
demanding than for a HF (3-30MHz) receiver.
Extending the published 1296MHz SSB transceiver design
to 432MHz or even to 144MHz does not make much sense. The
dynamic range of the subharmonic mixers used in the 1296MHz
RTX is certainly not sufficient for the current usage of the
144MHz amateur band. Due to several high-power stations, even
direct AM demodulation in the simple subharmonic mixers would
become a problem. A zero-IF SSB transceiver for 144MHz
requires much better mixers operating at the fundamental LO
frequency.
On the other hand it is not easy to find suitable
components for very high frequencies. Only components for
10GHz (transistors, diodes etc) can be readily found on the
market and it takes much effort to make them work on 24GHz.
There are few parts suitable for 47GHz or higher frequencies.
In addition, frequency accuracy, mixer balance and quadrature,
mechanical stability and efficient shielding are more
difficult to handle at higher frequencies.
In this article a zero-IF SSB transceiver for 10368MHz
will be described. The design is based on the same components
as the 5760MHz transceiver: HEMTs are used as amplifiers and
BAT14-099R schottky quads are used as subharmonic mixers.
Although 10GHz does not represent a problem for HEMTs yet,
this seems to be the upper limit for the BAT14-099R quads.
The BAT14-099R quads are packaged in the relatively large and
unsymmetric SOT-143 package causing severe mixer unbalance
problems.
The 10GHz SSB transceiver also includes PIN-diode antenna
switching. The RF front-end is built on teflon laminate, while
all other microstrip circuits are built on conventional FR4
glassfiber-epoxy, including several bandpass filters for 10GHz.
The IF and AF sections are of course identical to those used
in the transceivers for 1296, 2304 and 5760MHz.
2. Modified VCXO and multiplier stages
Although the radio-amateur 10GHz frequency-band allocation
extends from 10000MHz to 10500MHz, most narrowband operation
is concentrated slightly above 10368MHz. In the near future
one may also expect narrowband activity in the satellite
segment (10450-10500MHz), probably concentrated around the
lower end around 10450MHz.
10368MHz is an integer multiple of many popular
frequencies. For example, 10368MHz is the ninth harmonic of
1152MHz, a reference frequency that also generates 2304MHz,
3456MHz, 5760MHz and 24192MHz. 10368MHz is also the eighth
harmonic of 1296MHz suggesting the use of the same VCXO and
multiplier chain for 648MHz with some additional multiplier
stages (X8) to obtain 5184MHz for the subharmonic mixers.
However, it makes sense to modify the VCXO itself for
operation in the 10GHz transceiver. The relative crystal
frequency pulling should be eight times smaller at 10GHz due
to the additional multiplier stages. On the other hand,
frequency stability is much more critical at 10368MHz than it
is at 1296MHz.
Both requirements can be met by replacing the original
VCXO using a 18MHz fundamental-resonance crystal with a
modified VCXO using a 27MHz third-overtone crystal. Overtone
crystals have a higher Q than fundamental crystals therefore
providing better frequency stability. On the other hand,
the frequency-pulling range of an overtone crystal is very
restricted and it is barely sufficient for a 10GHz
transceiver.
Fortunately the circuit diagrams of both VCXOs are
similar and the same printed-circuit board can be used for
both of them, including the multiplier chain up to 648MHz.
Of course the 10GHz transceiver requires an additional
multiplier for 5184MHz built in microstrip technology just
like in the 5760MHz transceiver. The 648MHz input frequency
is first multiplied by four to 2592MHz and then doubled to
5184MHz.
The modified VCXO and multiplier chain up to 648MHz are
shown on Fig.1. A 27.000MHz crystal is required for
operation at 10368MHz. The frequency-pulling range is very
small in spite of the large capacitance ratio of the MV1404
varactor. The frequency coverage amounts to only 150-200kHz
at the final frequency centered around 10368.100MHz.
A coverage of 150kHz is however sufficient for normal
10GHz narrowband work, provided that the frequency stability
is reasonable. Fortunately the 27.000MHz crystal is used in
teletext decoders inside many TV sets. Thanks to very high
volume production these inexpensive crystals have an excellent
frequency stability.
The overtone oscillator itself (BFX89) is designed to
reduce the loading and therefore the heating of the crystal,
to further improve the frequency stability. The VCXO is
followed by two resonant circuits (L2 and L3) tuned to 54MHz.
The following multiplier stages are identical to those used
in the 1296MHz transceiver and are tuned to 162MHz (X3),
324MHz (X2) and 648MHz (X2).
Since the VCXO module is followed by an additional
multiplier for 5184MHz in the 10368MHz transceiver, an output
power of 10mW (+10dBm) is sufficient at 648MHz. Therefore the
power-supply resistors of the multiplier stages inside the
VCXO module are increased to 330ohm, 330ohm and 220ohm. Since
the printed-circuit board is the same as in the 1296MHz
version and there are just a few minor variations in the
component location, the corresponding drawings will not be
published once again.
The circuit diagram of the additional multiplier for
5184MHz is shown on Fig.2. The circuit includes four HEMTs
ATF35376. The first HEMT is overdriven by the 648MHz signal
to produce many harmonics. The following microstrip bandpass
(L3, L4, L5, L6, L7 and L8) selects the fourth harmonic at
2592MHz. The second HEMT amplifies the 2592MHz signal to
drive the third HEMT operating as a frequency doubler. The
doubler is followed by a 5184MHz bandpass (L15, L16, L17, L18,
L19 and L20) and the 5184MHz signal is finally amplified by
the last HEMT to obtain 20mW (+13dBm).
The additional multiplier for 5184MHz is built on a
double-sided microstrip 0.8mm FR4 board with the dimensions of
20mmX120mm as shown on Fig.3. The corresponding component
location is shown on Fig.4. The 5184MHz multiplier should
provide the rated output power (+13dBm) without any tuning,
provided that all of the components are installed and
grounded correctly.
3. Quadrature transmit mixer for 10368MHz
The circuit diagram of the quadrature transmit mixer for
10368MHz is shown on Fig.5. The 5184MHz LO signal is taken from
a -15dB coupler and the LO signal level is restored by the
ATF35376 amplifier stage, feeding two subharmonic mixers
equipped with BAT14-099R schottky quads. The 5GHz lowpass
attenuates the second harmonic at 10GHz to avoid corrupting
the symmetry of the mixers.
The unwanted carrier rejection of the BAT14-099R
subharmonic mixers is only 10-15dB at 10368MHz. The main
reason for the rather poor carrier rejection is the large and
unsymmetrical SOT-143 package. Pin 1 of this SMD package is
wider than the remaining three pins. Unfortunately, microwave
schottky quads in suitable symmetric packages are hardly
available and much more expensive than the BAT14-099R.
On the other hand, the subharmonic mixer balance can be
corrected by a DC bias applied to the diodes. The quadrature
transmit mixer for 10368MHz therefore includes two 10kohm
trimmers to adjust the mixer balance. In this way the unwanted
carrier rejection can be improved to better than 30dB.
The two 10GHz signals are combined in a quadrature
hybrid. The hybrid is built as a 100ohm circuit to save board
space. Quarter-wave transformers are used to restore the
imepdances to 50ohms. The circuit diagrams of both subharmonic
mixers are slightly different from those used in the 1296,
2304 or 5760MHz versions for the same reason.
The hybrid is followed by a 10368MHz bandpass filter
(L36, L37, L38, L39 and L40). The latter removes the 5184MHz
LO as well as other unwanted mixing products. After filtering
the 10368MHz SSB signal level is rather low (around 30uW or
-15dBm), so two amplifier stages with ATF35376 HEMTs are used
to boost the output signal level to about 2.5mW (+4dBm).
The quadrature transmit mixer for 10368MHz is built on a
double-sided microstrip 0.8mm FR4 board with the dimensions of
30mmX120mm as shown on Fig.6. The corresponding component
location is shown on Fig.7. Since only positive bias is
available through the 10kohm trimmers for mixer balancing,
teh BAT14-099R packages should be oriented correctly in the
circuit. In particular, one of the two packages has to be
installed upside down.
Except for the balancing trimmers, the quadrature transmit
mixer for 10368MHz should not require any tuning. Both
balancing trimmers are simply adjusted for the minimum output
power when no modulation is present. This adjustment is best
performed when the whole transceiver is assembled.
4. RF front-end for 10368MHz
The circuit diagram of the RF front-end for 10368MHz is
shown on Fig.8. The RF front-end includes a transmit power
amplifier, a receive low-noise amplifier and a PIN-diode
antenna switch. Unlike the RF front-ends for 1296, 2304 or
5760MHz, all built on FR4 laminate, the RF front-end for
10368MHz is built as a microstrip circuit on 0.5mm thick
teflon laminate to reduce RF losses.
Building the microstrip circuit on glassfiber-teflon
laminate (thickness 0.5mm or 0.020", dielectric constant
Er=2.55) allows an 1-2dB increase of the transmitter output
power and an 1-2dB improvement of the receiver noise figure.
The output power of the 10368MHz transmitter is in fact
even slightly higher than the output power of the 5760MHz
transmitter. Although both transmitters use the same
semiconductor devices (HEMTs), the latter is built on lossy
FR4 laminate.
The transmitter power amplifier is designed with
inexpensive HEMTs, so the output power is limited to 100mW
(+20dBm) on the antenna connector. The amplifier includes
an ATF35376 HEMT driver stage followed by two ATF35376 HEMTs
in parallel in the output stage. The 100ohm resistor between
L5 and L6 improves the power dividing and prevents push-pull
parasitic oscillations of the output stage.
The two output HEMTs receive a positive bias on the gates
both while transmitting and while receiving. During
transmission the transistors generate a self bias just like
in the 5760MHz transmitter. Of course the +4VTX supply line
requires a current-limiting resistor.
The antenna switch is using a single shunt PIN diode just
like the 5760MHz counterpart with the PIN diode BAR80.
However, the parasitic capacitance of the BAR80 is far too
high for operation at 10368MHz, so the new diode BAR81 (SMD
component marking "ABs" or "BBs", same MW-4 package) has
to be used. The parasitic capacitance of the new BAR81 PIN
diode is less than half that of the old BAR80. The insertion
loss of the BAR81 is reduced by applying a negative bias
(+-PIN) while receiving.
During transmission, the BAR81 is turned on and the short
circuit is transformed by L15 into an open circuit at the
summing node. The insertion loss of the BAR81 in the receive
path exceeds 20dB and this is sufficient to protect the
receiver. During reception, the two transmitter output HEMTs
act as short circuits thanks to the positive gate bias. The
short circuit is transformed through package parasitics, L7,
L8 and interconnecting lines (total electrical length 3/4
lambda) into an open circuit at the summing node.
Since there are no strong signals expected in the 10GHz
band, the LNA for 10368MHz includes two stages with ATF35376
HEMTs. The total insertion gain including the losses in the
antenna switching network and the two 10368MHz bandpass
filters amounts to about 23dB.
The RF front-end for 10368MHz is built on a double-sided
microstrip 0.5mm glassfiber-teflon board with the dimensions
of 30mmX80mm as shown on Fig.9. The corresponding component
location is shown on Fig.10. A low-loss teflon laminate
allows a higher output power and a better noise figure.
On the other hand, a low-loss teflon laminate does not
suppress the parasitic oscillations of HEMTs in the millimeter
frequency range. These oscillations have to be controlled by
damping resistors, usually 100ohm connected between gate and
source.
The RF frnt-end for 10368MHz includes a single tuning
point. A capacitive tuning stub (a 2mmX3mm piece of copper
foil) is added to L13 to improve the antenna matching,
which in turn depends on the installation of the antenna cable
and connector. This matching stub will improve the output
power by about 1dB or in other words, the output power may
be as low as 80mW without any tuning.
5. Quadrature receive mixer for 10368MHz
The circuit diagram of the quadrature receive mixer for
10368MHz is shown on Fig.11. It differs from the similar
5760MHz mixer in the design of the quadrature hybrid and
subharmonic mixers with BAT14-099R diodes. The module includes
two RF amplifiers with ATF35376 HEMTs, two 10368MHz bandpass
filters, two subharmonic mixers operating in quadrature and
two identical IF preamplifiers with BF199 transistors.
The subharmonic mixers are identical to those in the
transmitter using BAT14-099R schottky quads. Both mixers are
supplied in phase (L49 and L50) with the LO signal. The RF
input signal is split by an 100ohm quadrature hybrid (L25,
L26, L27 and L28). Impedance matching is provided by
quarter-wavelength lines L23, L29 and L30.
The mixers are followed by two IF preamplifiers with
BF199 transistors identical to those used in the 1296, 2304
and 5760MHz receivers. The 3.3mH chokes can be replaced by
lower values, since powerful signals (both amateur and
out-of-band) are not expected in the 10GHz band. Lower value
chokes are less sensitive to disturbing low-frequency magnetic
fields, including disturbing fields generated in the
transceiver itself.
The quadrature receiving mixer for 10368MHz is built on a
double-sided microstrip 0.8mm FR4 board with the dimensions of
30mmX120mm as shown on Fig.12. The corresponding component
location is shown on Fig.13. In the 10GHz band, a quarter
wavelength is only 4mm long on FR4 or 5mm long on teflon
boards, so more complicated microstrip circuits can be used.
For example, the supply/bias chokes are built as two-section
lowpass filters on the RF-front-end teflon board and as
three-section lowpass filters on both mixer FR4 boards. These
improved RF chokes introduce less insertion loss and allow a
a lower crosstalk.
The receiving mixer for 10368MHz requires no tuning.
However, the BF199 IF preamplifier transistors should be
selected for the lowest noise. It seems that these transistors
do not have a guaranteed "1/f" noise specification. In all of
the prototypes built the Philips BF199 transistors produced
the least amount of noise.
6. Construction of the zero-IF SSB transceiver for 10368MHz
The SSB/CW transceiver for 10368MHz is using the same
quadrature modulator, IF amplifier and demodulator as the
similar transceivers for 1296, 2304/2320 and 5760MHz. Since
the same semiconductors are used as in the 5760MHz version,
the current-limiting resistor in the SSB/CW switching RX/TX
module has to be set to 82ohm, 1W.
The new PIN diode BAR81 also requires the small PIN
driver module also used in the 5760MHz transceiver. Since
the new BAR81 is much improved with respect to the old BAR80,
the resulting receiver sensitivity loss amounts only to a few
dB if no negative bias is provided (no PIN driver used) and
the +-PIN line is simply connected to +12VTX. It therefore
makes sense to use the new BAR81 also in the 5760MHz
transceiver, while the old BAR80 is good enough for 1296 or
2304/2320MHz.
In the 10GHz frequency range, even small SMD components
are rather large when compared to the wavelength of only 29mm.
The size of the SMD resistors and capacitors is usually
indicated in hundredths of an inch (0.254mm). The first SMD
resistors and capacitors were of the size 1206 (about
3mmX1.5mm). Today, most SMD components are available in the
sizes 0805 and 0603, while the newest components are of the
size 0402 (dimensions 1mmX0.5mm).
Large 1206 SMD capacitors should not be used in the
10368MHz transceiver, since they have parasitic internal
resonances in the 10GHz frequency range. The resonant
frequencies are further decreased by the high dielectric
constant of the ceramic used to build these capacitors. In the
10368MHz transceiver only 0805 or smaller SMD parts should
be used.
In the 10GHz frequency range it makes sense to use
small-value capacitors (mainly 6.8pF in the circuit diagrams),
since they are built from low-loss WHITE ceramic with a
moderate dielectric constant and internal resonances above
18GHz. Higher-value capacitors made from coloured ceramic
(purple or brown) have higher RF losses and lower resonance
frequencies. Finally, the newest and smallest 0402 resistors
and capacitors are useful even at 24GHz.
The main building block of the RF circuits operating at
10368MHz are the ATF35376 HEMTs, although there are many
similar devices produced by other manufacturers that offer the
same S-parameters at similar bias conditions. When selecting
these devices one should take care of the Idss, since most
of these transistors operate at zero bias for circuit
simplicity.
An Idss of about 30mA is desirable. Devices with higher
Idss are only useful in the transmitter output amplifier. If
devices with a sufficiently low Idss can not be obtained, then
the 270-ohm 1/2W (or similar) resistors should be reduced to
allow proper operation of the zener or LED shunt regulator.
Lower NF selections of the same device, like the ATF35176 or
the ATF35076, usually have a higher Idss!
The intermediate/audio frequency section of zero-IF or
direct-conversion transceivers also requires a careful
selection of active devices with low "1/f" of "popcorn" noise.
Experience accumulated by building many transceivers for 1296,
2304/2320, 5760 and 10368MHz shows that Philips BF199
transistors perform best in the IF preamplifiers. However,
several very noisy samples of the BC238 transistors had to be
replaced in the quadrature IF amplifiers as well. It seems that
factory rejects fairly exceeding the "1/f" noise requirements
are sent to hobbyst shops. On the other hand, industrial
leftovers found at fla markets perform best, since these
devices had to go through severe input quality controls.
The RF modules of the 10368MHz SSB transceiver have the
same dimensions as the corresponding units of the 5760MHz
counterpart, but the exact locations of the electrical
connections are slightly different. Their shielded enclosures
should be carefully manufactured out of thin brass sheet,
since the 10368MHz SSB transceiver is even more sensitive to
microphonics and RF leakage. Both transmit and receive mixers
and the RF front-end require 1cm thick microwave absorber foam
(antistatic foam) installed under the covers of the shielded
enclosures.
The VCXO module does not require a bottom cover like in
the 1296MHz version, since 10368MHz is only the sixteenth
harmonic of 648MHz. On the other hand, the additional
multiplier for 5184MHz and the RF modules should be accurately
shielded. Efficient feedthrough capacitors should be used
elsewhere. All internal RF connections should be made with
double-braid flexible teflon coax or UT085 semirigid.
The complete 10368MHz SSB transceiver can be installed in
the same enclosure as the 1296, 2304 or 5760MHz versions with
a central chassis and internal dimensions of 60mm (height) X
180mm (width) X 180mm (depth). The module location within
the enclosure is the same as in the 5760MHz version, including
a similar wiring among the modules.
The loudspeaker should not be installed in the same case
to avoid microphonics. Microphonics could perhaps be reduced
by using machined enclosures for the RF modules. Mechanical
vibrations can also be controlled by inserting pieces of
plastic foam between the modules to act as an acoustic
absorber.
7. Checkout of the zero-IF SSB transceiver for 10368MHz
The checkout of the transceiver should start with the
alignment of the VCXO and multiplier stages. The VCXO should
be adjusted for the desired frequency coverage. The multiplier
stages are simply adjusted for the maximum output at the
desired frequency. The maximum is observed as the rectified
voltage drop on the base of the following transistor, measured
through a suitable RF choke. The maximum on 648MHz is measured
as the drain-current dip of the 2592MHz multiplier. Although
the additional multiplier does not require any adjustments,
the output signal level (+13dBm at 5184MHz) should be checked.
Since the receiver does not require any tuning, it should
already work. First, the overall amplification should be
checked. The output noise should drop when the supply to the
LNA is removed. When the IF preamplifiers are disconnected,
the noise should drop almost to zero.
Next the receiver is connected to an antenna and tuned to
a weak unmodulated carrier (distant beacon etc). Besides the
desired signal its rather weak image should also be heard in
the loudspeaker. The image can be detected since its frequency
changes in the opposite direction when tuning the receiver.
The image is then attenuated by adjusting the two trimmers,
phase quadrature and amplitude balance, in the IF amplifier.
The transmitter should be first checked for the output
power. The full output power should be acheived with the
trimmer in the modulator at about 1/3 resistance in CW mode.
The DC voltage across the PA transistors should rise to the
full voltage allowed by the 4V7 zener. Finally, the output
power is optimised with the tuning stub on L13.
The transmitter is then switched to SSB to adjust the
balance of both transmit mixers. The two 10kohm trimmers are
simply adjusted for the minimum output power with no
modulation. Unfortunately this setting is sensitive to the
mixer-diode temperature and 5184MHz LO drive level, so the
carrier suppression may not stay as good as it was adjusted
during the checkout of the transceiver.
The SSB modulation should be checked in a radio contact
with another amateur station on 10368MHz. In particular the
correct sideband, USB or LSB, should be checked, since the
I and Q modulation lines are easily interchanged by mistake.
The other station should also check the carrier leakage or
transmit mixer unbalance, heard as a 1365Hz tone added to
the modulation.
Finally, the shielding of the transceiver should be
checked. Waving your hand in front of the antenna usually
causes a 1365Hz whistle in the loudspeaker of the receiver.
The latter is caused by local oscillator leakage, frequnecy
shifted by the Doppler effect of the moving hand and finally
collected by the antenna. While this effect can not be
eliminated completely with the suggested mechanical
construction, it should be small enough to allow normal use
of the transceiver. Of course, the receiver sensitivity and
shielding can also be checked with a mains-operated
fluorescent tube as described for the 1296, 2304 and 5760MHz
transceivers.
The current drain of the described 10368MHz transceiver
should be around 390mA during quiet reception at a nominal
supply voltage of 12.6V. The current drain of the transmitter
is inversely proportional to the output power and ranges
from 550mA (CW or SSB peak power) up to 580mA (SSB no
modulation). The current drain could be reduced substantially
if a more efficient regulator were used to power the several
HEMT stages with an operating voltage of only 2V.
[1] Matjaz Vidmar: "NO-TUNE SSB TRANSCEIVERS FOR 1296, 2304 & 5760MHz".
List of figures:
Fig. 1 - VCXO and multipliers for 648MHz.
Fig. 2 - Additional multiplier for 5184MHz.
Fig. 3 - 5184MHz multiplier PCB (0.8mm double-sided FR4).
Fig. 4 - 5184MHz multiplier component location.
Fig. 5 - Quadrature transmit mixer for 10368MHz.
Fig. 6 - 10368MHz TX mixer PCB (0.8mm double-sided FR4).
Fig. 7 - 10368MHz TX mixer component location.
Fig. 8 - RF front-end for 10368MHz.
Fig. 9 - 10368MHz RF front-end PCB (0.5mm glassfiber-teflon Er=2.55).
Fig. 10 - 10368MHz RF front-end component location.
Fig. 11 - Quadrature receive mixer for 10368MHz.
Fig. 12 - 10368MHz RX mixer PCB (0.8mm double-sided FR4).
Fig. 13 - 10368MHz RX mixer component location.
