NO-TUNE SSB/CW TRANSCEIVER FOR
1296, 2304 AND 5760MHz
Matjaz Vidmar, S53MV
1. Microwave SSB transceiver design
When discussing SSB transceivers, the first question to
be answered is probably the following: does it make sense to
develop and build new SSB radios? Today SSB transceivers are
mass-produced items for frequencies below 30MHz. There is
much less choice on the market for 144MHz or 432MHz SSB
transceivers and there are just a few products available for
1296MHz or even higher frequencies.
Most radio-amateurs are therefore using a base SSB
transceiver (usually a commercial product) operating on a lower
frequency and suitable receive and transmit converters or
transverters to operate on 1296MHz or higher frequencies.
The most popular base transceiver is certainly the good old
IC202. All narrow-band (SSB/CW) microwave activity is therefore
concentrated in the first 200kHz of amateur microwave segments
like 1296.000-1296.200, 2304.000-2304.200 etc due to the
limited frequency coverage of the IC202.
Transverters should always be considered a poor technical
solution for many reasons. Receive converters usually degrade
the dynamic range of the receiver while transmit converters
dissipate most of the RF power generated in the base SSB
transceiver. Both receive and transmit converters generate
a number of spurious mixing products that are very difficult
to filter out due to the harmonic relationships among the
amateur frequency bands 144/432/1296...
However, the worst problem of most transverters is
the breakthrough of strong signals in or out of the
base-transceiver intermediate-frequency band. This problem
seems to be worst when using a 144MHz first IF. Strong 144MHz
stations with big antenna arrays may break in the first IF even
at distances of 50 or 100km. Since the problem is reciprocal, a
careless microwave operator may even establish two-way contacts
on 144MHz although using a transverter and antenna for 1296MHz
or higher frequencies.
Some microwave operators solved the above problem by
installing a different crystal in the transverter, so that
for example 1296.000MHz is converted to a less used segment
around 144.700MHz. Serious microwave contestmen use
transverters with a first IF of 28MHz, 50MHz or even 70MHz to
avoid the abovementioned problem. Neither solution is cheap.
The biggest problem is to carry a large 144MHz or HF all-mode
transceiver together with a suitable power supply on a
mountaintop.
Even the good old IC202 has its own problems. This radio
is no longer being manufactured for more than a decade. New
radios can not be purchased while the maintenance of the old
ones is becoming difficult. Second-hand radios are usually
found in very poor conditions due to the many "modifications"
and "improvements" made by their previous owners.
As a conclusion, today it still makes sense to develop
and build SSB radios for 1296MHz and higher frequencies. Since
the abovementioned problems of the transverters are well known
and are not really new, many technical solutions were
considered by different designers. Most solutions were
discarded simply because too complex, too expensive and too
difficult to build, even when compared to the already complex
combination of a base RTX and transverter.
Most commercial SSB transceivers include a modulator
and a demodulator operating on a high IF, as shown on Fig.1.
The resulting SSB signal is converted to the RF operating
frequency in the transmitter and back to the IF in the
receiver. Both the transmitter and the receiver use expensive
components like crystal filters. Besides crystal filters,
additional filtering is required in the RF section to
attenuate image responses and spurious products of both
receiving and transmitting mixers.
The design of conventional (high-IF) SSB transceivers
dates back to the vacuum-tube age, when active components
(tubes) were expensive and unreliable. Passive components
like filters were not so critical. Complicated tuning
procedures only represented a small fraction of the overall
cost of a vacuum-tube SSB transceiver.
SSB crystal filters usually operate in the frequency
range around 10MHz. A double or even triple upconversion is
required to reach microwave frequencies in the transmitter.
On the other hand, a double or triple downconversion is
required in the receiver to get back to the crystal-filter
frequency. Commercial VHF/UHF SSB transceivers therefore save
some expensive components by sharing some stages between the
transmitter and the receiver.
A conventional microwave SSB transceiver is therefore
complicated and expensive. Building such a transceiver in
amateur conditions is difficult at best. Lots of work as well
as some microwave test equipment is required. The final result
is certainly not cheaper and may not perform better than the
well-known transverter + base RTX combination.
Fortunately, expensive crystal filters and complicated
conversions are not essential components of a SSB transceiver.
There are other SSB transceiver designs that are both cheaper
and easier to build in amateur conditions. The most popular
seems to be the direct-conversion SSB transceiver design shown
on Fig.2. A direct-conversion SSB receiver acheives most of
its gain in a simple audio-frequency amplifier, while the
selectivity is acheived by simple RC lowpass filters.
The most important feature of a direct-conversion SSB
transceiver is that there are no complicated conversions nor
image frequencies to be filtered out. The RF section of a
direct-conversion SSB transceiver only requires simple LC
filters to attenuate far-away spurious responses like
harmonics and subharmonics. In a well designed
direct-conversion SSB transceiver, the RF section may not
require any tuning at all.
The most important drawback of a direct-conversion SSB
transceiver is a rather poor unwanted sideband rejection.
The transmitter includes two identical mixers operating at
90 degrees phase shift (quadrature mixer) to obtain only one
sideband. The receiver also includes two identical mixers
operating at 90 degrees phase shift to receive just one
sideband and suppress the other sideband. A direct-conversion
SSB transceiver operates correctly only if the gain of both
mixers is the same and the phase shift is exactly 90 degrees.
A direct-conversion SSB transceiver therefore includes
some critical components like precision (1%) resistors,
precision (2%) capacitors, selected or "paired" semiconductors
in the mixers and complicated phase-shifting networks. The
most complicated part is usually the audio-frequency 90-degree
divider or combiner including several operational amplifiers,
precision resistors and capacitors. Although using precision
components, the unwanted sideband rejection will seldom be
better than -40dB. This is certainly not enough for serious
work on HF.
In spite of the abovementioned difficulties,
direct-conversion designs are quite popular among the builders
of QRP HF transceivers. At frequencies above 30MHz it is
increasingly more difficult to obtain accurate phase shifts.
Due to the low natural (antenna) noise above 30MHz, a low-noise
RF amplifier is usually used to improve the mixer noise figure.
A LNA may cause direct AM detection in the mixers. A LNA may
also corrupt the amplitude balance and phase offset of the
two mixers, if the local oscillator signal is picked up by the
antenna. A VHF direct-conversion SSB transceiver is therefore
not as simple as its HF counterpart.
On the other hand, a direct-conversion SSB design has
important advantages over conventional SSB transceivers with
crystal filters, since there are no image frequencies and less
spurious responses. Professional (military) SSB transceivers
therefore use direct conversion, but the AF phase shifts are
obtained by digital signal processing. The DSP uses an adaptive
algorithm to measure and compensate any errors like amplitude
unbalance or phase offset of the two mixers, to obtain a
perfect unwanted sideband rejection.
Additional AF signal processing also allows a different
SSB transceiver design, for example a SSB transceiver with
a zero IF as shown on Fig.3. The latter is very similar to a
direct-conversion RTX except that the local oscillator is
operating in the center of the SSB signal spectrum, in other
words at an offset of about 1.4kHz with respect to the
SSB suppressed carrier frequency.
In a zero-IF SSB transceiver, the audio frequency band
from 200Hz to 2600Hz is converted in two bands from 0 to
1200Hz. Lowpass filters therefore have a cutoff frequency of
1200Hz, thus allowing a high rejection of the unwanted
sideband. A zero-IF SSB transceiver therefore retains all of
the advantages of a direct-conversion design and solves the
problem of the unwanted-sideband rejection.
The quadrature IF amplifier of a zero-IF SSB transceiver
includes two conventional AF amplifiers. Since the latter are
usually AC coupled, the missing DC component will be converted
in the demodulator in a hole in the AF reponse around 1.4kHz.
Fortunately this hole is not harmful at all for voice
communications, since it coincides with a hole in the spectrum
of the human voice. In fact, some voice-communication
equipment includes notch filters to create an artificial
hole around 1.4kHz to improve the signal-to-noise ratio and/or
to add a low-baud-rate-telemetry channel to the voice channel.
Therefore a potential drawback of a zero-IF design is actually
an advantage for voice communications.
Like a direct-conversion RTX, a zero-IF SSB transceiver
also requires quadrature transmit and receive mixers. However,
amplitude unbalance or phase errors are much less critical,
since they only cause distortion of the recovered audio signal.
Conventional components, like 5% resistors, 10% capacitors and
unselected semiconductors may be used anywhere in a zero-IF
SSB transceiver.
Finally, a zero-IF SSB transceiver does not require
complicated phase-shifting networks. Both the quadrature
modulator in the transmitter and the quadrature demodulator
in the receiver (phasor rotation and counterrotation with
1.4kHz) are made by simple rotating switches and fixed
resistor/opamp networks. CMOS analog switches like the 4051
are ideal for this purpose, rotated by digital signals coming
from a 1.4kHz oscillator.
Although the block diagram of a zero-IF SSB transceiver
looks complicated, such a transceiver is relatively easy to
build. In particular, very little (if any) tuning is required,
since there are no critical components used anywhere in the
transceiver. In particular, the RF section only includes
relatively wideband (10%) bandpass filters that require no
tuning. The IF/AF section also accepts wide component
tolerances and thus requires no tuning. The only remaining
circuit is the RF local oscillator. The latter may need some
tuning to bring the radio to the desired operating frequency...
2. Microwave SSB transceiver implementation
The described zero-IF concept should allow the design
of simple and efficient SSB transceivers for an arbitrary
frequency band. Three successful designs of zero-IF SSB
transceivers covering the lower amateur microwave bands of
1296MHz, 2304MHz and 5760MHz will be described in this
article. Similar technical solutions were first tested in PSK
packet-radio transceivers operating at 1.2Mbit/s in the
23cm and 13cm amateur frequency bands.
Of course several requirements and technology issues
need to be considered before a theoretical concept can
materialize in a real-world transceiver. Fortunately the
requirements are not severe for the lower amateur microwave
bands. In this frequency range no very strong signals are
expected, so there are no special requirements on the dynamic
range of the receiver. Only a relatively limited frequency
range needs to be covered (200 to 400kHz in each band) and
this can be easily acheived using a VXO and multipliers as
the local oscillator.
From the technology point of view it is certainly
convenient to use up-to-date components. High-performance and
inexpensive microwave semiconductors were developed first
for satellite-TV receivers and then for mobile communications
like GSM or DECT telephones. These new devices provide up to
25dB of gain per stage up to 2.3GHz and up to 14dB of gain per
stage up to 10GHz. Many other functions, like schottky mixer
diodes or antenna-switching PIN diodes are also available.
Using obsolete components makes designs complicated.
For example, the well known transistors BFR34A and BFR91
were introduced almost 25 years ago. At that time they were
great devices providing almost 5dB of gain at 2.3GHz. Today
it makes more sense to use an INA-03184 MMIC to get 25dB
of gain at 2.3GHz or in other words replace a chain of 5 (five)
amplifier stages with the abovementioned obsolete transistors.
The availability of active components also influences the
selection of passive components. Many years ago, all microwave
circuits were built in waveguide technology. Waveguides allow
very low circuit losses and high-Q resonators. Semiconductor
microwave devices introduced microstrip circuits built on
low-loss substrates like alumina (Al2O3) ceramic or
glassfiber-teflon laminates. Conventional glassfiber-epoxy
laminates like FR4 were not used above 2GHz due to the high
losses and poor Q of microstrip resonators.
However, a zero-IF SSB transceiver design does not require
a very high selectivity in the RF section. If the circuit
losses can be compensated by high-gain semiconductor devices,
cheaper substrates like the conventional glassfiber-epoxy FR4
can be used at frequencies up to at least 10GHz. The FR4
laminate has excellent mechanical properties. Unlike soft
teflon laminates, cutting, drilling and hole plating in FR4
is well known. Even more important, most SMD component packages
are designed for installation on a FR4 substrate and may break
or develop intermittent contacts if installed on a soft teflon
board.
Therefore, losses in FR4 microstrip transmission lines and
filters were investigated. Surprisingly, the losses were found
inversely proportional to board thickness and rather slowly
increasing with frequency. This simply means that the FR4 RF
losses are mainly copper losses, while dielectric losses are
still rather low. FR4 RF copper losses are high since the
copper surface is made very rough to ensure good mechanical
bonding to the dielectric substrate.
In fact, if the copper foil is peeled off a piece of FR4
laminate, the lower foil surface is rather dark. On the other
hand, if the copper foil is peeled off a piece of microwave
teflon laminate, the colours of both foil surfaces are similar.
Since different manufacturers use different methods for bonding
the copper foil, RF losses are different in different FR4
laminates. On the other hand, the dielectric constant of FR4
was found quite stable. Finally, silver or gold plating of
microstrip lines etched on FR4 laminate really makes no sense,
since most of the RF losses are caused by the (inaccessible)
rough foil surface bonded to the dielectric.
A practical FR4 laminate thickness for microwave circuits
with SMD components is probably 0.8mm. A 50-ohm microstrip line
has a width of about 1.5mm and about 0.2dB/cm of loss at
5.76GHz. Therefore microstrip lines have to be kept short if
etched on FR4 laminate. For comparison, the FR4 microstrip
losses are about three times larger than the microstrip losses
of a glassfiber-teflon board and about ten times larger than
the losses of teflon semirigid coax cables.
Although FR4 laminate losses are high, resonators and
filters can still be implemented as microstrip circuits.
Considering PCB etching tolerances and especially underetching,
both transmission lines and gaps in between them should not
be made to narrow. A practical lower limit is 0.4mm width for
the transmission lines and 0.3mm for the gaps.
A practical 5.76GHz two-resonator bandpass design is
shown on Fig.4. The measured insertion loss of 3.5dB is
referred to the worst-case, very lossy FR4. A better FR4 could
get down to 3dB or even 2.5dB. Although an insertion loss of
3.5dB is rather high for a 10% bandwidth filter, it can easily
be recovered with modern high-gain semiconductor devices.
For comparison, the insertion loss of a SMD coupling capacitor
may be as high as 0.5dB.
As already mentioned, modern semiconductor devices are
really easy to use even at microwave frequencies. Silicon MMIC
amplifiers provide 25dB of gain (limited by package parasitics)
up to 2.3GHz. If less gain is required, conventional silicon
bipolar transistors can be used, since their input and output
impedances are also close to 50ohms.
GaAs semiconductors are more practical above about 5GHz.
In particular, high performance devices like HEMTs became
inexpensive since they are mass produced for satellite-TV
receivers. HEMTs operate at lower voltages and higher currents
than conventional GaAsFETs, so their input and output
impedance are very close to 50ohms at frequencies above 5GHz.
Serious microwave engineers are afraid of using HEMTs
since these devices have enough gain to oscillate at
frequencies above 50GHz or even 100GHz. In this case it is
actually an advantage to build the circuit on a lossy laminate
like FR4, since the latter will efficiently suppress any
oscillations in the millimeter frequency range. Having the
ability to control the loss in a circuit therefore may
represent an advantage!
The availability of inexpensive power GaAsFETs greatly
simplifies the construction of transmitter output stages.
In particular, the high gain of power GaAsFETs in the 23cm
and 13cm bands greatly reduces the number of stages when
compared to silicon bipolar solutions.
Zero-IF and direct-conversion transceivers have some
additional requirements for mixers. Mixer balancing is very
important, both to suppress the unwanted residual carrier in
the transmitter and to suppress the unwanted AM detection in
the receiver. At microwave frequencies, the simpliest way of
acheiving good mixer balancing is to use a subharmonic mixer
with two antiparallel diodes as shown on Fig.5.
Such a mixer requires a local oscillator at half
frequency. Frequency doubling is acheived internally in the
mixer circuit. A disadvantage of this mixer is a higher noise
figure in the range 10 to 15dB and sensitivity to the LO
signal level. Both a too-low LO drive or a too-high LO drive
will further increase the mixer insertion loss and noise
figure.
On the other hand, the abovementioned subharmonic mixer
only requires two non-critical microstrip resonators that
do not influence the balancing of the mixer. The best
performances were obtained using schottky quads with the
four diodes internally connected in a ring. The schottky
quad BAT14-099R provides about -35dB of carrier suppression
at 1296MHz and about -25dB of carrier suppression at 5760MHz
with no tuning.
A very important advantage of the subharmonic mixer is
that the local oscillator operates at half of the RF frequency.
This reduces the RF-LO crosstalk and therefore the shielding
requirements in zero-IF or direct-conversion transceivers.
A side advantage is that the half-frequency LO chain requires
less multiplier stages.
The three zero-IF SSB transceivers for 1296MHz, 2304MHz
and 5760MHz have many parts in common. In particular, the
AF and IF sections are identical in all three transceivers.
The RF sections are similar, however the microstrip filters
are necessarily different as well as the low-noise and
power devices used in each frequency band. Finally, the same
VCXO module is used, with small modifications, in all three
transceivers.
Therefore the individual modules will be described
first. Of course, similar modules for different frequency
ranges will be described together. Finally, an overview
of the construction techniques of the individual modules
will be given, as well as shielding of the modules and
integration of the complete transceivers.
3. VCXO and multipliers
Since a relatively narrow frequency range needs to be
covered, a VXO followed by multiplier stages is an efficient
solution for the local oscillator. The VXO is built as a
varactor-tuned VCXO with a fundamental-resonance crystal,
since the frequency-pulling range of overtone crystals is not
sufficient for this application. A fundamental-resonance
crystal has a lower Q and is less stable than overtone
crystals, but for this application the performance is
sufficient.
Fundamental resonance crystals can be manufactured for
frequencies up to about 25MHz. Therefore the output of the
VCXO needs to be multiplied to obtain microwave frequencies.
Frequency multiplication can be obtained by a chain of
conventional multipliers including class-C amplifiers and
bandpass filters or by a phase-locked loop.
Although the PLL requires almost no tuning and is
easily reproducible, the PLL solution was discarded for other
reasons. A SSB transceiver requires a very clean LO signal,
therefore the PLL requires buffer stages to avoid pulling
the VCXO and/or the microwave VCO. Shielding and power-supply
regulation are also critical, making the whole PLL multiplier
more complicated than a conventional multiplier chain.
The circuit diagram of the VCXO and multiplier stages
is shown on Fig.6. The VCXO is operating around 18MHz in
the transceivers for 1296MHz and 2304MHz and around 20MHz
in the transceiver for 5760MHz. All multiplier stages use
silicon bipolar transistors BFX89 (BFY90) except the last
stage with a BFR91. The module already supplies the required
frequency of 648MHz for the 1296MHz version of the transceiver.
In the 2304MHz version, the module supplies 576MHz by
using different multiplication factors. The latter frequency
is doubled to 1152MHz inside the transmit and receive mixer
modules. In the 5760MHz version, the module supplies 720MHz
and this frequency is further multiplied to 2880MHz in an
additional multiplier module. Of course, the values of a few
components need to be adjusted according to the exact
operating frequency, shown in () brackets for 2304MHz and in
[] brackets for 5760MHz.
The VCXO and multiplier chain are built on a single-sided
FR4 board with the dimensions of 40mmX120mm as shown on Fig.7.
The corresponding component location (for the 648MHz version)
is shown on Fig.8. The exact value of L1 depends on the crystal
used. Some parallel-resonance crystals may even require
replacing L1 with a capacitor. L2 and L3 have about 150nH
each or 4 turns of 0.25mm copper-enamelled wire on a 10X10mm
IF-transformer coilformer. L4 and L5 are self-supporting
coils of 4 turns of 1mm copper-enamelled wire each, wound on
an internal diameter of 4mm. L6, L7, L8 and L9 are etched on
the PCB.
The VCXO module is the only part of the whole transceiver
that requires tuning. L2, L3 and the capacitors in parallel
with L4, L5, L6, L7, L8 and L9 should simply be tuned for
the maximum output at the desired frequencies. In a multiplier
chain, RF signal levels can easily be checked by measuring
the DC voltages over the BE junctions of the multiplier
transistors.
When the multiplier chain is providing the specified
output power, L1 and the capacitor in parallel with the
MV1404 varactor should be set for the desired frequency
coverage of the VCXO. If standard "computer grade" 18.000MHz
or 20.000MHz crystals are used, it is recommended to select the
crystal with the smallest temperature coefficient.
Unfortunately not all amateurs are allowed to use the
international segment around 2304MHz on 13cm. It is a little
bit more difficult to find a crystal for 18.125MHz for the
German segment around 2320MHz.
The 5760MHz transceiver requires an additional multiplier
from 720MHz to 2880MHz as shown on Fig.9. The first HEMT
ATF35376 operates as a quadrupler while the second HEMT
ATF35376 operates as a selective amplifier for the output
frequency of 2880MHz. The additional multiplier for 2880MHz
is built on a double-sided microstrip FR4 board with the
dimensions of 20mmX120mm as shown on Fig.10. The corresponding
component location is shown on Fig.11.
The 2880MHz multiplier should provide the rated output
power of +11dBm without any tuning. On the other hand, the
tuning of L8 and L9 to 720MHz in the VCXO module can be
optimized for the minimum DC drain current (max DC voltage)
of the first HEMT. The two red LEDs are used as 2V zeners.
LEDs are in fact better than real zeners, since they have a
sharper knee and do not produce any avalanche noise.
4. SSB/CW quadrature modulator
The main purpose of the SSB/CW quadrature modulator is
to convert the input audio frequency band from 200Hz to 2600Hz
into two bands 0 to 1200Hz to drive the quadrature transmit
mixer. Additionally the module includes a microphone amplifier
and a circuit to generate the CW signal. The circuit diagram
of the modulator module is shown on Fig.12.
The microphone amplifier includes two stages with the
transistors BC238. The input is matched to a low-impedance
dynamic mike with the 33ohm resistor. The 1N4007 diode protects
the input in the case the microphone input is simply connected
in parallel to the loudspeaker output. Finally the output
drives an emitter follower with another BC238.
The CW carrier is generated in the same way as the SSB
transmission. The 683Hz square wave, coming from the
demodulator module, is first cleaned in a low-pass audio filter
and then processed in the same way as a SSB signal. Both AF
modulation sources are simply switched by 1N4148 diodes.
The main component of the modulator is the 4051 CMOS
analog switch. The switch is rotated with the 1365Hz, 2731Hz
and 5461Hz clocks coming from the demodulator. The input audio
signal is alternatively fed to the I and Q chains. The I and Q
signals are obtained with a resistor network and the first
four opamps (first MC3403). Then both I and Q signals go
through lowpass filters to remove unwanted mixing products.
Finally there are two voltage followers to drive the
quadrature transmit mixer.
The SSB/CW quadrature modulator is built on a single-sided
FR4 board with the dimensions of 40mmX120mm as shown on Fig.13.
The corresponding component location is shown on Fig.14. Most
components are installed vertically to save board space.
The SSB/CW quadrature modulator does not require any alignment.
The 4.7kohm trimmer is provided to check the overall
transmitter. Full power (in CW mode) should be obtained with
the trimmer cursor in central position.
5. Quadrature transmit mixers
All three transmit mixer modules for 1296MHz, 2304MHz
and 5760MHz include similar stages: a LO signal switching,
an in-phase LO divider, two balanced subharmonic mixers, a
quadrature combiner and a selective RF amplifier. LO signal
switching between the transmit and receive mixers is performed
in the following way: most of the LO signal is always fed to
the receive mixer. A small fraction of the LO signal is
obtained from a coupler and amplified to drive the transmit
mixer. During reception the power supply of the LO amplifier
stage is simply turned off. This solution may look
complicated, but in practice it allows an excellent isolation
between the transmit and receive mixers. The practical circuit
is simple and the component count is low as well.
The circuit diagram of the quadrature transmit mixer for
1296MHz is shown on Fig.15. The 648MHz LO signal is taken from
a -20dB coupler and the LO signal level is restored by
the BFP183 amplifier stage, feeding two subharmonic mixers
equipped with BAT14-099R schottky quads. The 648MHz lowpass
attenuates the second harmonic at 1296MHz to avoid corrupting
the symmetry of the mixers.
The two 1296MHz signals are combined in a quadrature
hybrid, followed by a 1296MHz bandpass filter. The latter
removes the 648MHz LO as well as other unwanted mixing
products. After filtering the 1296MHz SSB signal level is
rather low (around -10dBm), so an INA-10386 MMIC is used to
boost the output signal level to about +15dBm.
The quadrature tansmit mixer for 1296MHz is built on a
double-sided microstrip FR4 board with the dimensions of
40mmX120mm as shown on Fig.16. The corresponding component
location is shown on Fig.17. The circuit does not require any
tuning for operation at 1296MHz or 1270MHz.
The circuit diagram of the quadrature transmit mixer for
2304MHz is shown on Fig.18. The 576MHz LO signal is taken from
a -20dB coupler, amplified by the BFP183 transistor and
then doubled to 1152MHz by the BFP196 transistor. The doubler
output goes through a microstrip bandapss filter to feed
the two subharmonic mixers with BAT14-099R schottky quads.
The two 2304MHz signals are combined in a quadrature
hybrid, followed by a 2304MHz bandpass filter. The latter
removes the 1152MHz LO as well as other unwanted mixing
products. After filtering the 2304MHz SSB signal level is
rather low (around -11dBm), so an INA-10386 MMIC is used to
boost the output signal level to about +10dBm.
The quadrature transmit mixer for 2304MHz is built on a
double-sided microstrip FR4 board with the dimensions of
40mmX120mm as shown on Fig.19. The corresponding component
location is shown on Fig.20. The circuit does not require any
tuning for operation at 2304MHz or 2320MHz. For operation in
the satellite band above 2400MHz, the LO bandpass should be
readjusted to 1200MHz by shortening L7 and L8 at their hot
ends.
The circuit diagram of the quadrature transmit mixer for
5760MHz is shown on Fig.21. The 2880MHz LO signal is taken from
a -15dB coupler and the LO signal level is restored by the
ATF35376 amplifier stage, feeding two subharmonic mixers
equipped with BAT14-099R schottky quads. The 2880MHz lowpass
attenuates the second harmonic at 5760MHz to avoid corrupting
the symmetry of the mixers.
The two 5760MHz signals are combined in a quadrature
hybrid, followed by a 5760MHz bandpass filter. The latter
removes the 2880MHz LO as well as other unwanted mixing
products. After filtering the 5760MHz SSB signal level is
rather low (around -14dBm), so two amplifier stages with
ATF35376 HEMTs are used to boost the output signal level to
about +11dBm.
The quadrature transmit mixer for 5760MHz is built on a
double-sided microstrip FR4 board with the dimensions of
30mmX120mm as shown on Fig.22. The corresponding component
location is shown on Fig.23. The circuit does not require any
tuning for operation at 5760MHz.
The quadrature transmit mixers do not supply any output
signal when the modulation input is absent. For transmiter
testing purposes it is possible to obtain an output signal
by feeding a DC current of 2-10mA into one or both mixers.
6. RF front-ends
The RF front-ends include the transmitter power
amplifiers, the receiver low-noise amplifiers and the antenna
switching circuits. Of course there are major differences
among different power amplifier designs, depending not just
on the frequency, but also on the technology used and the
output power desired. On the other hand, it no longer makes
sense to use expensive coaxial relays, since PIN diodes
can provide the same insertion loss and isolation at lower
cost with better reliability and much shorter switching times.
The circuit diagram of the RF front-end for 1296MHz is
shown on Fig.24. The transmitter power amplifier includes a
single stage with a CLY5 power GaAsFET, providing a gain of
15dB and an output power of about 1W (+30dBm). The CLY5 is a
low-voltage transistor operating at about 5V.
The negative gate bias is generated by rectification of
the driving RF signal in the GS junction inside the CLY5
during modulation peaks. The gate is then held negative for
a few seconds thanks to the 1uF storage capacitor. To prevent
overheating and destruction of the CLY5, the +5VTX voltage
is obtained through a current-limiting resistor. This
arrangement may look strange, but it is very simple, requires
no adjustments, allows a reasonably linear operation and most
important of all, it proved very reliable in PSK packet-radio
transceivers operating 24 hours per day in our packet-radio
network.
The antenna switch includes a series diode BAR63-03W and
a shunt diode BAR80. Both diodes are turned on while
transmitting. L9 is a quarter-wavelength line that transforms
the BAR80 short circuit into an open for the transmitter.
The receiving preamplifier includes a single BFP181 transistor
(15dB gain) followed by a 1296MHz bandpass filter (-3dB loss).
In the 1296MHz RF front-end, the LNA gain should be limited
to avoid interference from powerful non-amateur users of this
band (radars and other radionavigation aids).
The RF front-end for 1296MHz is built on a double-sided
microstrip FR4 board with the dimensions of 40mmX80mm as shown
on Fig.25. The corresponding component location is shown on
Fig.26. The RF front end for 1296MHz requires no tuning.
However, since the output impedance of the INA-10386 inside
the transmit mixer is not exactly 50ohms, the cable length
between the transmit mixer and the RF front-end is critical.
Therefore L1 may need adjustments if the teflon-dielectric
cable length is different from 12.5cm.
The circuit diagram of the RF front-end for 2304MHz is
shown on Fig.27. The transmitter power amplifier includes
two stages: a BFP183 driver and a CLY2 final amplifier.
The additional BFP183 driver is required since the gain and
output power of the INA-10386 inside the transmit mixer are
smaller at 2304MHz than at 1296MHz. Further the additional
bandpass filter at the input of the power amplifier adds some
insertion loss.
The BFP183 operates as a class-A amplifier while the
CLY2 is used in a similar self-biasing arrangement like the
CLY5 in the 1296MHz front-end. Of course the drain current of
the CLY2 is smaller, the +5VTX current-limiting resistor must
be higher and the RF output power on the antenna connector
amounts to about 0.5W (+27dBm). The PIN-diode antenna switch
is identical to that used in the 1296MHz RF front-end with a
series diode BAR63-03W and a shunt diode BAR80.
Since there are no powerful users of the 2.3GHz band,
the RF front-end for 2304MHz includes a two-stage LNA:
a HEMT ATF35376 in the first stage and a BFP181 in the second
stage. The overall gain of the LNA is around 23dB. Since the
Idss of the ATF35376 is usually around 30mA, no negative
voltage needs to be applied to the gate.
The RF front-end for 2304MHz is built on a double-sided
microstrip FR4 board with the dimensions of 40mmX80mm as shown
on Fig.28. The corresponding component location is shown on
Fig.29. The RF front-end for 2304MHz should require no
tuning, since both the transmit and the receive chains have
a few dB of gain margin. However, in order to squeeze the
last milliwatt out of the CLY2 (is it really necessary?),
some tuning may be attempted on the output.
The circuit diagram of the RF front-end for 5760MHz is
shown on Fig.30. The transmit power amplifier uses two HEMTs
ATF35376 in parallel to obtain about 100mW (+20dBm) on the
antenna connector. The gain of the HEMTs is around 13dB,
however circuit losses both in the input matching network
and in the antenna switch on the output amount to about 3dB,
so that about +10dBm of drive power is required.
The two PA HEMTs receive a positive bias on the gates both
while transmitting and while receiving. In transmission the PA
HEMTs generate a self bias just like the CLY5 and CLY2 power
GaAsFETs. Of course the +4VTX supply line requires a
current-limiting resistor.
The antenna switch includes a single shunt diode BAR80
to protect the receiver input during transmission. During
reception, the two PA HEMTs act as short circuits thanks to
the positive gate bias. The short circuit is transformed
through package parasitics, L5, L6 and interconnecting lines
(total electrical length 3/4 lambda) into an open circuit
at the summing node.
Since the shunt diode BAR80 was not designed for operation
above 3GHz, its capacitance introduces additional insertion
loss in the receiving path at 5.76GHz. This additional loss can
be substantially reduced if a reverse bias is applied to the
BAR80 diode. Therefore a negative bias voltage is applied
to the BAR80 during reception and a positive current is
applied during transmission through the command line +-PIN.
Since there are no strong signals expected in the 5.7GHz
band, the LNA for 5760MHz includes two stages with ATF35376
HEMTs. The total insertion gain including the losses in
the antenna switching network and the two 5760MHz bandpass
filters amounts to about 23dB.
The RF front-end for 5760MHz is built on a double-sided
microstrip FR4 board with the dimensions of 30mmX80mm as shown
on Fig.31. The corresponding component location is shown on
Fig.32. The RF front-end for 5760MHz requires no tuning.
It is however recommended to select the ATF35376 HEMTs
according to the Idss. The highest Idss devices should be
used in the transmitter PA while the lowest Idss devices
should be used in the receiver LNA.
7. Quadrature receive mixers
All three receiving mixer modules for 1296MHz, 2304MHz
and 5760MHz include similar stages: an additional RF signal
amplifier, a quadrature-hybrid divider, two subharmonic
mixers, an in-phase LO divider and two IF preamplifiers.
The mixers, in-phase and quadrature dividers and RF bandpass
filters are very similar to those used in the transmitting
mixer modules.
The circuit diagram of the quadrature receiving mixer for
1296MHz is shown on Fig.33. The incoming RF signal is first
fed through a microstrip bandpass filter, then amplified
with an INA-03184 MMIC and further filtered by another,
identical microstrip bandpass. The total gain of the chain
of the two filters and the MMIC is about 20dB.
A high gain in the RF section is required to cover the
relatively high noise figure of the two subharmonic mixers
and the additional losses in the quadrature hybrid. The
two receiving subharmonic mixers are also using BAT14-099R
schottky quads. The mixer outputs are fed through lowpass
filters to the IF preamplifiers.
The IF preamplifiers are using HF transistors BF199.
These were found to perform better than their BC...
counterparts in spite of the very low frequencies involved
(less than 1200Hz). HF transistors have a smaller current
gain, their input impedance is therefore smaller and better
matches the output impedance of the mixers. Both IF
preamplifiers receive their supply voltages from the IF
amplifier module.
The quadrature receiving mixer for 1296MHz is built on a
double-sided microstrip FR4 board with the dimensions of
40mmX120mm as shown on Fig.34. The corresponding component
location as shown on Fig.35. The receiving mixer for 1296MHz
requires no tuning.
The circuit diagram of the quadrature receiving mixer for
2340MHz is shown on Fig.36. Since the same components,
INA-03184 MMIC and BAT14-099R schottky quads, have similar
performances in the 1296MHz and 2304MHz frequency bands, the
circuit diagram of the 2304MHz mixer is almost identical to
the 1296MHz mixer.
The only difference is the additional frequency doubler
to 1152MHz with the transistor BFP196. The multiplier includes
a lowpass on the input and a bandpass filter on the output.
The input lowpass filter should prevent unwanted interactions
with other circuits operating with the same 576MHz LO signal.
The 1uH choke should have a ferrite core for the same reason.
The quadrature receiving mixer for 2304MHz is built on a
double-sided microstrip FR4 board with the dimensions of
40mmX120mm as shown on Fig.37. The corresponding component
location is shown on Fig.38. The receiving mixer does not
require any tuning for operation on 2304MHz or 2320MHz. For
operation in the satellite band above 2400MHz, the LO bandpass
should be readjusted to 1200MHz by shortening L26 and L27 at
their hot ends.
The circuit diagram of the quadrature receiving mixer for
5760MHz is shown on Fig.39. Since a component with the gain
comparable to the INA-03184 is not available for 5.7GHz,
two RF amplifier stages are required to obtain about 20dB
of gain. ATF35376 HEMTs are used in both RF amplifier
stages. Otherwise the circuit is almost identical to the
quadrature receiving mixer for 1296MHz.
The quadrature receiving mixer for 5760MHz is built on a
double-sided microstrip FR4 board with the dimensions of
30mmX120mm as shown on Fig.40. The corresponding component
location is shown on Fig.41. The receiving mixer for 5760MHz
requires no tuning.
8. SSB zero-IF amplifier with AGC
The basic feature of direct-conversion and zero-IF
receivers is to acheive most of the signal gain with simple
and inexpensive AF amplifiers. Further, the selectivity is
acheived with simple RC lowpass filters that require no
tuning. The circuit diagram of such an IF amplifier equipped
with AGC is therefore necessarily different from conventional
high-IF amplifiers.
A zero-IF receiver requires a two-channel IF amplifier,
since both I and Q channels need to be amplified independently
before demodulation. The two IF channels should be as much
identical as possible to preserve the amplitude ratio and
phase offset between the I and Q signals. Therefore both
channels should have a common AGC so that the amplitude ratio
remains unchanged.
The circuit diagram of the quadrature SSB IF amplifier
with AGC is shown on Fig.42. The IF amplifier module includes
two identical lowpass filters on the input, followed by a
dual-amplifier stage with a common AGC. An amplitude/phase
correction is performed after the first amplifier stage,
followed by another pair of lowpass filters and another
dual-amplifier stage with a common AGC.
The two input lowpass filters are active RC filters using
BC238 emitter followers. Discrete bipolar transistors are used
because much less noisy than operational amplifiers. The input
circuit also provides the supply voltage to the IF
preamplifiers inside the receiving mixer module through the
1.5kohm resistors.
The dual-amplifier stages are also built with discrete
BC238 bipolar transistors. Each amplifier stage includes a
voltage amplifier (first BC238) followed by an emitter-follower
(second BC238) essentially to avoid mutual interactions when
the amplifiers are chained with other circuits in the IF strip.
The AGC is using MOS transistors as variable resistors
on the inputs of the dual-amplifier stages. To keep the gain
of both I and Q channels identical, both MOS transistors are
part of a single integrated circuit 4049UB. The digital CMOS
integrated circuit 4049UB is being used in a rather uncommon
way, however the remaining components inside the 4049UB act
just as diodes and do not disturb the operation of the AGC.
The IF amplifier module includes two trimmers for small
corrections of the amplitude balance (10kohm) and phase
offset (250kohm) between the two channels. The correction
stage is followed by two active RC lowpass filters using
operational amplifiers (MC3403), since the signals are already
large enough and the operational-amplifier noise is no longer
a problem. Finally there is another, identical dual-amplifier
stage with its own AGC.
The quadrature SSB IF amplifier is built on a single-sided
FR4 board with the dimensions of 50mmX120mm as shown on Fig.43.
The corresponding component location is shown on Fig.44. In
order to keep the differences between the I and Q channels
small, good quality componets should be used in the IF
amplifier. Using 5% resistors, 10% foil-type capacitors and
conventional BC238B transistors should keep the differences
between the two channels small enough for normal operation.
Most components are installed vertically to save board space.
The amplitude balance (10kohm) and phase offset (250kohm)
trimmers are initially set to their neutral (central) position.
These trimmers are only used while testing the complete
receiver to obtain the minimum distortion of the reproduced
audio signal.
9. Quadrature SSB demodulator and AF amplifier
The main function of the quadrature SSB demodulator is
the conversion of both I and Q IF signals (frequency range
0 to 1200Hz) back to the original audio frequency range from
200Hz to 2600Hz. The same module includes a power AF amplifier
and a clock generator for both the phasor rotation in the
transmitter and the phasor counterrotation in the receiver.
The circuit diagram of the module is shown on Fig.45.
The quadrature SSB demodulator includes four operational
amplifiers (MC3403) to produce an 8-phase system from the
I and Q signals, using a resistor network similar to that
used in the modulator. The signal demodulation or phasor
counterrotation is performed by the CMOS analog switch 4051,
rotating with a frequency of 1365Hz. The I and Q signals are
alternatively fed to the output or in other words the circuit
performs exactly the opposite operation of the modulator.
Unwanted mixing products of the phasor counterrotation
are removed by an active RC lowpass (BC238). The demodulated
audio signal is fed to the 100kohm volume control. A LM386 is
used as the audio power amplifier due to its low current drain
and small external component count.
The three clocks required to rotate both 4051 switches
in the modulator and in the demodulator are supplied by a
binary counter 4029. The 4029 includes an up/down input that
allows the generation/demodulation of USB or LSB in this
application. The up/down input has a 100kohm pullup resistor
for USB operation. LSB is obtained when the up/down input
is grounded through a front-panel switch.
USB/LSB switching is usually not required for terrestrial
microwave work. USB/LSB switching is only required when
operating through satellites or terrestrial linear
transponders or when using inverting converters or
transverters for other frequency bands. Finally, USB/LSB
switching may be useful to attenuate interferences during CW
reception. An alternative way of USB/LSB switching is
interchanging the I and Q channels. When assembling together
the modules of the described transceivers one should therefore
check the wiring so that the transmitter and the receiver
operate on the same sideband at the same time.
The 4029 counter requires an input clock around 11kHz.
This clock does not need to be particularly stable and a RC
oscillator could be sufficient. In the described transceivers
a crystal source was preferred to avoid any tuning. In
addition, if all transceivers use the same rotation or
counterrotation frequency, the mutual interferences are
reduced.
The crystal oscillator is using a clock crystal operating
on a relatively low frequency of 32768Hz. The dual D-flip-flop
4013 divides this frequency by 3 to obtain a 10923Hz clock
for the 4029 binary counter. The resulting rotation frequency
for the 4051 switches is 1365Hz. The latter figure almost
perfectly matches the hole in the frequency spectrum of human
voice. The same 4029 counter also supplies the CW tone 683Hz
to reduce the unwanted mixing products in the transmitter.
The quadrature SSB demodulator and AF amplifier are built
on a single-sided FR4 board with the dimensions of 40mmX120mm
as shown on Fig.46. The corresponding component location is
shown on Fig.47. Most of the components are installed
vertically to save board space.
The 32768Hz crystal oscillator will only operate
reliably with a 4011UB (or 4001UB) integrated circuit. The
commonly available "B" series CMOS integrated circuits
(4011B in this case) have a too-high gain for this
application. In the latter case a 560pF capacitor may help
to stabilize the oscillator. On the other hand, the oscilator
circuit usually works reliably with old 4011 or 4001 circuits
with an "A" suffix or no suffix letter at all.
10. SSB/CW switching RX/TX
A SSB/CW transceiver requires different switching
functions. Fortunately both SSB and CW modes of operation
require the same functions in the receiver. Of course two
different operating modes are required for the transmitter:
SSB voice and CW keying.
The RX/TX changeover is controlled by the PTT switch
on the microphone in the SSB mode of operation. In the CW
mode of operation, most transceivers use an automatic delay
circuit to keep the transmitter enabled during CW keying.
This delay circuit was perhaps required in old radios using
several mechanical relays. In modern transceivers with all
electronic switching it makes no sense, since the RX/TX
switching can be performed in less than one millisecond.
It therefore makes sense that SSB transmission is enabled
by simply pressing the PTT switch while CW transmission is
enabled by pressing the CW key so that no special (and useless)
controls are required on the front panel of the trasceiver.
In the CW mode of operation, no delays are required and
the receiver is enabled immediately after the CW key is
released ("BK" mode of operation).
The circuit diagram of the SSB/CW switching RX/TX is
shown on Fig.48. In the described SSB/CW transceivers, most
modules are enabled at all times with a continuous +12V
supply: VCXO and multipliers, receiving mixer, IF amplifier,
demodulator and modulator. When enabling the transmitter
either by pressing the PTT or CW keys, the RX LNA is turned
off (+12VRX) and the TX PA is turned on (+12VTX and +5VTX
or +4VTX).
During SSB transmission the RX AF amplifier is turned
off (+12VAF), to avoid disturbing the microphone amplifier
(+12VSSB). On the other hand, during CW transmission the
AF amplifier remains on as well as most receiver stages, so
that CW keying can be monitored in the loudspeaker or phones.
The +12VCW supply connects the 683Hz signal to the modulator
input.
The supply voltages +12VAF, +12VSSB, +12VCW and +12VRX
are switched by BC327 PNP transistors. Due to the higher
current drain, the +12VTX supply voltage requires a more
powerful PNP transistor BD138. The TX PA receives its supply
voltage through a current-limiting resistor from the +12VTX
line. Since the latter dissipates a considerable amount of
power, it is built from several smaller resistors and located
in the switching unit to prevent heating the PA transistor(s).
The value of the current-limiting resistor depends on the
version of the transceiver. The 1296MHz PA with a CLY5 requires
8 pieces of 33ohm 1/2W resistors for a total value of 16.5ohms.
The 2304MHz PA with a CLY2 requires 4 pieces of 33ohm 1/2W
resistors for a total value of 33ohms. Finally, the 5760MHz PA
with two ATF35376s requires a single 82ohm 1W resistor.
The switching module also includes the circuits to drive
the front-panel meter. The latter is a moving-coil type with
a full-scale sensitivity of about 300uA. The meter has two
functions. During reception it is used to check the battery
voltage. The 8V2 zener extends the full scale of the meter
to the interesting range from about 9V to about 15V.
During transmission the meter is used to check the
supply voltage of the PA transistor(s). Due to the self-biasing
operation, the PA voltage will be only 0.5-1V without
modulation and will rise to its full value, limited by the
zener diode inside the PA, only when full drive is applied.
The operation of the PA and the output RF power level can
therefore be simply estimated from the PA voltage.
A S-meter is probably totally useless in small portable
transceivers as those described in this article. If desired,
the AGC voltge can be amplified and brought to a front-panel
meter. However, one should not forget that LED indicators are
not visible in full sunshine on a mountaintop, so the choice
is limited to moving-coil and LCD meters.
Most components of the SSB/CW switching RX/TX are
installed on a single-sided FR4 board with the dimensions
of 30mmX80mm as shown on Fig.49. Their location (1296MHz
version) is shown on Fig.50. Only the reverse-polarity
protection diode 1N5401 and the 470uF electrolytic capacitor
are installed directly on the 12V supply connector. The 10kohm
trimmer is used to adjust the meter sensitivity.
The 5760MHz version of the SSB/CW transceiver requires
an additional PIN diode driver to provide a negative bias
to the BAR80 PIN diode during reception. The corresponding
circuit diagram is shown on Fig.51. The negative voltage
is obtained from the 5461Hz clock, while the BC327 PNP
transistor applies a positive voltage during transmission.
The PIN driver is built on a small FR4 board with the
dimensions of 23mmX20mm as shown on Fig.52. The corresponding
component location is shown on Fig.53. The whole PIN driver
module is then installed piggy-back on the demodulator board,
using as support the 5-pole connector carrying the clocks to
the modulator.
11. Construction of zero-IF SSB transceivers
The described zero-IF SSB transceivers are using many
SMD parts in the RF section. SMD resistors usually do not
cause any problems, since they have low parasitics up to
at least 10GHz. On the other hand, there are big differences
among SMD capacitors. For this reason a single value (47pF)
was used anywhere. The 47pF capacitors used in the prototypes
are NPO type, rather large (size 1206), have a self-resonance
around 10GHz and introduce an insertion loss of about 0.5dB
at 5.76GHz. Finally, the 4.7uF SMD tantalum capacitors can be
replaced by the more popular tantalum "drops".
Quarter-wavelength chokes are used elsewhere in the RF
circuits. In the 5760MHz transceiver all quarter-wavelength
chokes are made as high-impedance microstrips. On the other
hand, to save board space in the 1296MHz and 2304MHz versions,
the quarter-wavelength chokes are made as small coils of 0.25mm
thick copper-enamelled wire of the correct length, chosen
according to the frequency: 12cm for 648MHz, 9cm for 23cm
mixers (648/1296MHz), 7cm for 1296MHz and for L3 on Fig.18,
5.5cm for 13cm mixers (1152/2304MHz) and 4cm for 2304MHz.
The wire is tinned for about 5mm on each end and the remaining
length is wound on an internal diameter of about 1mm.
The SMD semiconductor packages and pinouts are shown
on Fig.54. Please note that due to lack of space, the SMD
semiconductor markings are different from their type names.
Only the relatively large CLY5 transistor in a SOT-223
package has enough space to carry the full marking "CLY5".
The remaining components only carry one-, two- or three-letter
marking codes.
All of the microstrip circuits are built on double-sided
0.8mm thick FR4 glassfiber-epoxy laminate. Only the top side
is shown in this article, since the bottom side is left
unetched to act as a groundplane for the microstrips.
The copper surface should not be tinned nor silver or gold
plated. The copper-foil thickness should be preferably 35um.
Since the microstrip boards are not designed for
plated-through holes, care should be taken to ground all
components properly. Microstrip lines are grounded using
0.6mm thick silver plated wire (RG214 central conductors)
inserted in 1mm diameter holes at the marked positions and
soldered on both sides to the copper foil.
Resistors and semiconductors are grounded through 2mm,
3.2mm and 5mm diameter holes at the marked positions. These
holes are first covered on the groundplane side with pieces
of thin copper foil (0.1mm). Then the holes are filled with
solder and finally the SMD component is soldered in the
circuit.
Feedtrough capacitors are also installed in 3.2mm diameter
holes in the microstrip boards and soldered to the groundplane.
Feedthrough capacitors are used for supply voltages and
low-frequency signals. Some components like bias resistors,
zeners and electrolytic capacitors are installed on the
bottom side (as shown on the component-location drawings) and
connected to the feedthrough capacitors.
The VCXO/multiplier module and all of the microstrip
circuits are installed in shielded enclosures as shown on
Fig.55. Both the frame and the cover are made of 0.4mm thick
brass sheet. The printed-circuit board is soldered in the
frame at a height of about 10mm from the bottom. Additional
feedthrough capacitors are required in the brass walls.
RF signal connections are made using thin teflon-dielectric
coax like RG-188. The coax braid should be well soldered to
the brass frame all around the entrance hole. Finally the
frame is screwed on the chassis using sheet-metal screws.
The covers are kept in place thanks to the elastic brass
sheet, so they need not be soldered. An inspection of the
content is therefore possible at any time. The VCXO/multiplier
module is built on a single-sided board and therefore requires
both a top and a bottom cover. The remaining modules are all
built as microstrip circuits, so the microstrip groundplane
acts as the bottom cover and only the top cover is required.
The sizes and shapes of the microstrip circuit boards
are selected so that no resonances occur up to and including
2880MHz. Microwave absorber foam is therefore only required
in the three modules operating at 5760MHz. To avoid disturbing
the microstrip circuit, the microwave absorber foam is
installed just below the top cover.
The modules of a zero-IF SSB transceiver are installed in
a custom-made enclosure as shown on Fig.56. The most important
component is the chassis. The latter must be made of a single
piece of 1mm thick Al sheet to provide a common ground for
all modules. If a common ground is not available, the receiver
will probably self-oscillate in the from of "ringing" or
"whistling" in the loudspeaker, especially at higher volume
settings.
The chassis carries both the front and back panels as well
as the top and bottom covers. All of the connectors and
commands are available on the front panel. The later is screwed
to the chassis using the components installed: CW pushbutton,
SMA connector, meter and tuning helipot. The top and bottom
cover are made of 0.5mm thick Al sheet to save weight and are
screwed to the chassis using sheet-metal screws.
The shielded RF modules are installed on the top side of
the chassis where a height up to 32mm is allowed. The
audio-frequency bare printed-circuit boards are installed on
the bottom side of the chassis. The interconnections between
both sides of the chassis are made through five large-diameter
holes.
The location of the modules of the 1296MHz or 2304MHz
transceiver as well as the location of the connectors and
commands on the front panel are shown on Fig.57 for both sides
of the chassis. The 5760MHz transceiver has different RF
modules, so the corresponding module location is shown on
Fig.58. The location of the connectors and commands is the
same for all three transceivers.
The loudspeaker should not be installed inside the
transceiver, since the RF receiving modules are quite
sensitive to vibrations. On the other hand, the same
loudspeaker may also be used as a dynamic microphone for the
transmitter. The circuit is designed so that it allows a
simple parallel connection of the loudspeaker output and
the microphone input. The PTT and CW keys are simple switches
to ground.
12. Checkout of zero-IF SSB transceivers
The described SSB/CW transceivers for 1296MHz, 2304MHz
and 5760MHz do not require much tuning. The only module that
really requires tuning is the VCXO/multiplier. The latter is
simply tuned for the maximum output on the desired frequency.
Of course, the desired coverage of the VCXO has to be set with
a frequency counter.
After the VCXO/multiplier module is adjusted, the
remaining parts of the transceiver require a checkout to
locate defective components, soldering errors or insufficient
shielding. The receiver should already work and some noise
should be heard in the loudspeaker. The noise intensity should
drop when the power supply to the LNA is removed. The noise
should completely disappear when the receiving mixer module is
disconnected from the IF amplifier. A similar noise should be
heard if only one (I or Q) IF channel is connected.
Next the receiver is connected to an outdoor antenna far
away from the receiver and tuned to a weak unmodulated carrier
(a beacon transmitter or another VCXO/multiplier module at
a distance of a few ten meters). Tuning the receiver around
the unmodulated signal one should hear both the desired tone
and its much weaker mirror-image tone changing its frequency in
the opposite direction. The two trimmers in the IF amplifier
should be set so that the mirror tone disappears. The correct
function of the USB/LSB switch can also be checked.
Finally, the shielding of the receiver should be checked.
A small, handheld antenna (10-15dBi) is connected to the
receiver and the main beam of the antenna is directed into the
transceiver. If the noise coming from the loudspeaker changes,
the shielding of the local oscillator multiplier chain is
insufficient.
Next a mains-operated fluorescent tube (20W or 40W) is
turned on in the same room. A weak mains hum should only
be heard when the handheld antenna is pointed towards the
tube at 2-3m distance. If a clean hum without noise is heard
regardless of the antenna direction, the shielding of the
local oscillator multiplier chain is insufficient.
The transmitter should be essentially checked for the
output power. The full output power should be acheived with
the trimmer in the modulator in a middle position in CW mode.
The DC voltage across the PA transistor should rise to the
full value allowed by the 5.6V or 4.7V zener diode.
The output power should drop by an equal amount if only
I or only Q modulation is connected to the transmit mixer.
Finally the SSB modulation has to be checked with
another receiver for the same frequency band or best in a
contact with another amateur station at a distance of a
few km. This is the simpliest way to find out the correct
sideband, USB or LSB, of the transmitter, since the I and Q
channels can be easily interchanged by mistake in the wiring.
The residual carrier level of the transmit mixer should
also be checked. Due to the conversion principle this carrier
results in a 1365Hz tone in a correctly-tuned SSB receiver.
The carrier suppresion may range from -35dB in the 1296MHz
RTX down to only -20dB in the 5760MHz RTX. A poor carrier
suppression may be caused by a too high LO signal level or
by a careless installation of the BAT14-099R mixer diodes.
Note that the residual carrier can not be monitored on another
correctly-tuned zero-IF receiver, since it falls in the
AF response hole of the zero-IF receiver.
The current drain of the described transceivers should
be as follows. Receivers: 1296MHz:105mA, 2304MHz:175mA and
5760MHz:300mA. The current drain of the transmitters is
inversely proportional to the output power due to the
self-biasing of the PA. The minimum current drain corresponds
to SSB modulation peaks or CW transmission. Transmitters:
1296MHz:650-870mA, 2304MHz:490-640mA and 5760MHz:410-440mA.
All figures are given for a typical sample at a supply voltage
of 12.6V.
At the end, one should understand that zero-IF
transceivers also have some limitations. In particular, the
dynamic range of the receiver is limited by the direct AM
detection in the receiving mixer. If very strong signals are
expected, the LNA gain has to be reduced to avoid the above
problem. This is already done in the 1296MHz receiver, since
strong radar signals are quite common in the 23cm band. The
sensitivity to radar interference of the described zero-IF
1296MHz transceiver was found comparable to the conventional
transverter + 2mRTX combination.
On the other hand, the 2304MHz and 5760MHz transceivers
have a higher gain LNA. If the dynamic range needs to be
improved, the second LNA stage can simply be replaced with a
wire bridge in both transceivers. Of course, the internal LNA
gain has to be reduced or the LNA has to be completely
eliminated if an external LNA is used.
List of figures:
Fig. 1 - Conventional (high-IF) SSB transceiver design.
Fig. 2 - Direct-conversion SSB transceiver design.
Fig. 3 - Zero-IF SSB transceiver design.
Fig. 4 - 5.76GHz bandpass filter response.
Fig. 5 - Subharmonic mixer design.
Fig. 6 - VCXO and multipliers for 648MHz (576MHz) [720MHz].
Fig. 7 - VCXO PCB (0.8mm single-sided FR4).
Fig. 8 - VCXO component location (648MHz version).
Fig. 9 - Additional multiplier for 2880MHz.
Fig. 10 - 2880MHz multiplier PCB (0.8mm double-sided FR4).
Fig. 11 - 2880MHz multiplier component location.
Fig. 12 - SSB/CW quadrature modulator.
Fig. 13 - Modulator PCB (0.8mm single-sided FR4).
Fig. 14 - Modulator component location.
Fig. 15 - Quadrature transmit mixer for 1296MHz.
Fig. 16 - 1296MHz TX mixer PCB (0.8mm double-sided FR4).
Fig. 17 - 1296MHz TX mixer component location.
Fig. 18 - Quadrature transmit mixer for 2304MHz.
Fig. 19 - 2304MHz TX mixer PCB (0.8mm double-sided FR4).
Fig. 20 - 2304MHz TX mixer component location.
Fig. 21 - Quadrature transmit mixer for 5760MHz.
Fig. 22 - 5760MHz TX mixer PCB (0.8mm double-sided FR4).
Fig. 23 - 5760MHz TX mixer component location.
Fig. 24 - RF front-end for 1296MHz.
Fig. 25 - 1296MHz RF front-end PCB (0.8mm double-sided FR4).
Fig. 26 - 1296MHz RF front-end component location.
Fig. 27 - RF front-end for 2304MHz.
Fig. 28 - 2304MHz RF front-end PCB (0.8mm double-sided FR4).
Fig. 29 - 2304MHz RF front-end component location.
Fig. 30 - RF front-end for 5760MHz.
Fig. 31 - 5760MHz RF front-end PCB (0.8mm double-sided FR4).
Fig. 32 - 5760MHz RF front-end component location.
Fig. 33 - Quadrature receive mixer for 1296MHz.
Fig. 34 - 1296MHz RX mixer PCB (0.8mm double-sided FR4).
Fig. 35 - 1296MHz RX mixer component location.
Fig. 36 - Quadrature receive mixer for 2304MHz.
Fig. 37 - 2304MHz RX mixer PCB (0.8mm double-sided FR4).
Fig. 38 - 2304MHz RX mixer component location.
Fig. 39 - Quadrature receive mixer for 5760MHz.
Fig. 40 - 5760MHz RX mixer PCB (0.8mm double-sided FR4).
Fig. 41 - 5760MHz RX mixer component location.
Fig. 42 - Quadrature SSB IF amplifier with AGC.
Fig. 43 - IF amplifier PCB (0.8mm single-sided FR4).
Fig. 44 - IF amplifier component location.
Fig. 45 - Quadrature SSB demodulator and AF amplifier.
Fig. 46 - Demodulator PCB (0.8mm single-sided FR4).
Fig. 47 - Demodulator component location.
Fig. 48 - SSB/CW switching RX/TX, 1296MHz (2304MHz) [5760MHz].
Fig. 49 - SSB/CW switching RX/TX PCB (0.8mm single-sided FR4).
Fig. 50 - SSB/CW switching RX/TX component location (1296MHz).
Fig. 51 - PIN driver for 5760MHz.
Fig. 52 - PIN driver PCB (0.8mm single-sided FR4).
Fig. 53 - PIN driver component location.
Fig. 54 - SMD semiconductor packages and pinouts.
Fig. 55 - Shielded RF module enclosure.
Fig. 56 - Zero-IF SSB/CW transceiver enclosure.
Fig. 57 - 1296MHz (2304MHz) SSB/CW transceiver module location.
Fig. 58 - 5760MHz SSB/CW transceiver module location.
